This invention is in the field of data communications, and is more specifically directed to measurement of subscriber loop parameters from a central office location for Digital Subscriber Line (DSL) communications.
High-speed Internet access is now a widespread utility for many businesses, schools, and homes. As is fundamental in this art and as well known by most consumers, broadband communications according to Digital Subscriber Line (DSL) is a popular technology for providing high-speed communications over existing telephone lines, without disturbing conventional voice telephony (Plain Old Telephone Service, or “POTS”).
FIG. 1a illustrates a conventional subscriber loop carrying out DSL communications between server SVR at a central office (CO) and a subscriber workstation CWS at the client premises. In this example, these bidirectional communications are carried out over loop LOOP, which is implemented by conventional twisted pair wire. At the central office, CO DSL modem 10c is bidirectionally coupled by a bus on one side to network or Internet server SVR, and on another side by a connection to subscriber loop LOOP. Conversely, at the client premises, subscriber DSL modem 10s is bidirectionally coupled via a bus to client workstation CWS and bidirectionally coupled to subscriber loop LOOP. Each of DSL modems 10c, 10s include parallel-in-serial-out transmit function 12c, 12s, which performs the appropriate modulation of the data to be transmitted, and applies the modulated signal to front end/hybrid 15c, 15s, which is also in DSL modem 10c, 10s, respectively. DSL modems 10c, 10s also include serial-in-parallel-out receive function 14c, 14s, which demodulate received signals and produce the appropriate parallel data stream to the respective host SVR, CWS, at the central office and client premises, respectively. According to the popular asymmetric DSL (ADSL) technology, a frequency division multiplexing (FDM) scheme places the upstream and downstream communications in separate, non-overlapping, frequency bands.
Fundamentally, conventional ADSL communications is carried out by way of Discrete Multitone (DMT) modulation, which is a form of broadband communications. According to conventional DMT technology, the available spectrum is subdivided into many subchannels (e.g., 256 subchannels of 4.3125 kHz). Each subchannel is centered about a carrier frequency that is phase and amplitude modulated, typically by Quadrature Amplitude Modulation (QAM), in which each symbol value is represented by a point in the complex plane. As known in the art, other modulation techniques are used for other types of DSL communications.
The number of available symbol values for each subchannel, and thus the number of bits in each symbol communicated over that subchannel (i.e., the “bit loading”), is determined during initialization of the DMT communications session. The bit loading on each DMT subchannel depends upon the signal-to-noise ratio (SNR) of the loop at the subchannel frequency. For example, relatively noise-free and low attenuation subchannels may communicate data in ten-bit to fifteen-bit symbols, represented by a relatively dense QAM constellation with short distances between points in the constellation. On the other hand, noisy channels may be limited to only two or three bits per symbol, allowing a greater distance between adjacent points in the QAM constellation. Typically, the SNR of some subchannels is so poor that these subchannels are unloaded, carrying no bits. DMT modulation thus maximizes the data rate over each subchannel, permitting high speed access to be carried out even over relatively noisy and attenuated twisted-pair lines.
This bit loading process requires measurement of the transmission characteristics between the central office (CO) of the DSL service provider and the client premises, over the range of subchannel frequencies. The signal-to-noise ratio at each of the subchannel frequencies are conventionally determined at the start of each communications session, from which the bit loading is determined. Under conventional DSL standards, the bit loading is determined during the “training” sequence at the beginning of the session, and relies upon bidirectional communications between the DSL transceiver at the CO and the DSL transceiver at the customer premises.
In addition, knowledge of the transmission channel characteristics is also important in initially setting up DSL service at a customer premises. Under current technology and DSL infrastructure deployment, not all home and office locations are within the DSL signal range of a CO. It is therefore desirable to determine whether reasonable data rates to the premises of a potential customer can be achieved under nominal conditions. Because there is necessarily no DSL transceiver located at the customer premises prior to installation, the bidirectional communications techniques used in DSL “training” are not available. As such, measurement of the transmission characteristics between a CO and a possible customer premises must either be performed in a “single-ended” manner by measuring the characteristics from the CO end of the connection, or by deploying a service technician to the customer premises to participate in bidirectional communications with the CO. Of course, single-ended loop characterization from the CO is preferred, because it avoids a “truck roll”, to send a technician to the customer premises.
FIG. 1b illustrates the DSL subscriber loop for purposes of single-ended testing. In this approach, server SVR is bidirectionally coupled to DSL modem 10c at the central office, as before. Front-end/hybrid 15c drives signals on and receives signals from subscriber loop LOOP. However, as shown in FIG. 1b, there is no DSL modem at the end of loop LOOP at the customer premises. Rather, the termination of loop LOOP is considered simply as a load L, which is of course involved in the characterization of the transmission characteristics of loop LOOP. According to conventional single-ended loop test (SELT) methods, central office DSL modem 10c applies signals to loop LOOP, and characterizes the transmission parameters of loop LOOP based on the response of loop LOOP to these applied signals.
In modern DSL technology, three common SELT methods for measuring transmission line or loop characteristics are: load impedance measurements, frequency domain reflectometry (FDM), and time domain reflectometry (TDR). These conventional methods will now be described in connection with an example of conventional DSL modem line driver and receiver circuitry 18, which in combination with hybrid circuit 26 constitutes front-end hybrid circuit 15c, as illustrated in FIG. 2.
As shown in FIG. 2, conventional line driver and receiver circuitry 18c includes separate circuitry for the transmit and receive functions. On the transmit side, transmit path circuitry 20 receives signals from parallel-in-serial-out transmit function 12c, and it filters and otherwise processes these signals into the appropriate form for amplification and application to loop LOOP. The double-ended outputs of transmit path circuitry 20 are applied to amplifiers 22a, 22b, which amplify the modulated signal and apply the amplified signal to secondary windings of transformers 24a, 24b; the primary windings of transformers 24a, 24b are connected in series between the two conductors of transmission loop LOOP, according to the convention in the art. Transformers 24a, 24b may be implemented as a single physical transformer with split windings, as known in the art. Amplifiers 22a, 22b are conventional operational amplifiers, biased to amplify the output of transmit path 20 as appropriate. The passive components incorporated into line driver and receiver circuitry 18 for effecting such bias and amplifier control are present, in the conventional manner, and are not described here in detail. It is contemplated that those skilled in the art having reference to this specification will be readily able to design the amplifier circuitry and implement the appropriate biasing schemes, given the illustration of FIG. 2.
On the receive side, hybrid circuit 26 is coupled to the common side of the secondary windings of transformers 24a, 24b, and passes its received signals to receive path circuitry 28. Hybrid circuit 26 is a conventional isolation amplifier function, as used in conventional DSL modems, for isolating the receive path circuitry 28 from signals being transmitted by the transmit side of line driver and receiver circuitry 18, in the conventional manner. Receive path circuitry 28 filters and otherwise processes the received signals, and forwards these signals on to serial-in-parallel-out transmit function 14c. 
Active termination is included in conventional line driver and receiver circuitry 18, by way of amplifiers 30a, 30b. As is well known in the transmission line art, reflections in the signals applied to a transmission line can be avoided by matching the output impedance of the driver to the transmission line impedance. In the simple case, passive termination schemes simply inserts a matching impedance into the front-end circuitry. However, one-half the transmitted power of the signal is dissipated by such passive termination.
It is known to use active termination circuitry, which matches the loop impedance while dissipating much less power. In conventional front-end/hybrid circuit 15c, active termination is implemented by way of operational amplifiers 30a, 30b, which have inputs coupled to the secondary winding of transformers 24a, 24b, respectively. The outputs of amplifiers 30a, 30b are coupled to the feedback inputs of line drive amplifiers 12a, 12b. According to this conventional operation, a simulation factor of on the order of 1/11 can be achieved, meaning that impedance mismatching reflections are substantially eliminated, while only dissipating on the order of 10% of the power of the line driver circuitry.
In the arrangement of FIGS. 1b and 2, for purposes of SELT, several assumptions can be made. First, the output impedance of the line driver without considering the active termination circuit can be assumed to be zero. Second, the input impedance of hybrid circuit 26 can be assumed to be infinite. Third, a noise voltage Vn can be assumed to be zero mean stationary and ergodic.
For the impedance measurement method of SELT characterization of loop LOOP, with load L at the distal end of loop LOOP from modem 10c, the output voltage Vr at the output of receive path circuitry 28 can be considered, in the frequency domain, as a function of the transmit voltage Vt at the input to transmit path circuitry 20, as follows:Vr(ω)=ƒ1((ZL(ω))Ht(ω)Hr(ω)Hht(ω)Vt(ω)+ƒ2((ZL(ω))Hr(ω)Hhn(ω)Vn(ω)  (1)In this equation (1), the functions ƒ1(ZL(ω)) and ƒ2(ZL(ω)) are the ideal line driver trans fer functions. Transfer function Ht(ω) is the transfer function of transmit path circuitry 20, transfer function Hr(ω) is the transfer function of receive path circuitry 28, and transfer functions Hht(ω), Hhn(ω), are the transfer functions of hybrid circuit 26 responsive to echoed transmit voltage Vt and noise Vn, respectively.
Under the assumptions that noise Vn(ω) is zero mean stationary and ergodic, and that transmit voltage signal Vt(ω) is periodic, the noise term of the response equation for Vr(ω) can be removed by time-averaging the time-domain received voltage vr(i):
                                                                        v                _                            r                        ⁡                          (              i              )                                =                                    1              N                        ⁢                                          ∑                                  j                  =                  0                                                  N                  -                  1                                            ⁢                                                          ⁢                                                v                  r                                ⁡                                  (                                      i                    +                    Mj                                    )                                                                    ,                                  ⁢                              for            ⁢                                                  ⁢            i                    =          0                ,        1        ,        …        ⁢                                  ,                  M          -          1                                    (        2        )            where M is the period of vt, so that {overscore (v)}r depends only upon vt. This permits expression of the average received voltage signal, in the frequency domain, by:{overscore (V)}r(ω)=ƒ1((ZL(ω))Ht(ω)Hr(ω)Hht(ω)Vt(ω)   (3)With knowledge of the transfer functions Ht(ω), Hr(ω), Hht(ω), one can thus solve for the impedance ZL(ω) of transmission loop LOOP. This impedance measurement is typically executed by a minimization algorithm, by way of which the values of a set of loop physical parameters are iterated to minimize a cost function matching this expression of {overscore (V)}r. The cost function is typically the integral, over frequency, of the absolute value of the difference between a function of the set of loop physical parameters and the load impedance function ƒ1((ZL(ω)), where the load impedance is expressed in terms of a ratio of the received average voltage {overscore (V)}r(ω) and the product Ht(ω)Hr(ω)Hht(ω)Vt(ω). Least mean squares minimization is the typical minimization approach, following which an alignment algorithm may be applied to solve for the phase information of this impedance ZL(ω).
While this impedance measurement SELT approach is accurate for short transmission loops, it has been observed that this method is not particularly useful for longer transmission loops. At long loop lengths, the load impedance ZL is dominated by the characteristic impedance Z0 of loop LOOP, which is independent of the loop length and also independent of the termination type at the remote end (i.e., independently of load L of FIG. 1b.). It is therefore difficult to estimate the termination impedance from the determination of the overall loop impedance ZL. In addition, active termination circuitry, such as that implemented in the conventional arrangement of FIG. 2, has a frequency response that depends on the loop impedance itself. This makes measurement of the loop impedance ZL difficult when active termination is involved.
The other conventional SELT approaches are reflection-based, in that they rely on the partial reflection of a transmitted signal from locations along transmission loop LOOP at which impedances are mismatched. In the general sense, a reflection coefficient ρ is defined at an impedance mismatch location as:
                    ρ        =                                                            V                0                -                            ⁢                              ⅇ                                                      -                    γ                                    ⁢                                                                          ⁢                  D                                                                                    V                0                +                            ⁢                              ⅇ                                  γ                  ⁢                                                                          ⁢                  D                                                              =                                                    Z                1                            -                              Z                0                                                                    Z                1                            +                              Z                0                                                                        (        4        )            In this equation, the voltages V0+ and V0− are the nominal voltage amplitudes of the forward-going and reflected signals from the reflection location, respectively. Impedance Z0 is the characteristic impedance of the transmission line, while impedance Z1 is the mismatched impedance. Propagation constant γ=α+βi characterizes the transmission, and loop length D is the distance to the mismatch location. Following this expression, the short loop case (Z1=0) has a reflection coefficient ρ of −1, and the open loop case (with Z1 infinite) has a reflection coefficient ρ of +1. In general, the voltage and current at any point of the loop is the superposition of the forward and backward traveling voltage and current waveforms:V(x)=V0+e−γα+V0−eγα  (5a)I(x)=I0+e−γα+I0−eγα  (5b)where x is the position along the loop from the transmit end.
Two different reflectometry approaches for determining the characteristics of a DSL transmission loop are known. One approach, referred to as Frequency Domain Reflectometry (FDR), is based on the fundamental knowledge that, in the transmission loop context, a standing wave is formed at wavelengths for which the round trip distance between the signal source and the impedance mismatch location is a multiple; these wavelengths can easily be synchronously detected as voltage peaks in the spectrum. According to the FDR approach, the loop length D can be calculated from measurement of the characteristic resonant frequencies at which standing waves are present.
As is fundamental in this art, one can express the measured voltage V(x) at a point x along an unterminated lossless transmission fine as:V(x)=2V0+eβDt cos [β(x+D)]  (6)for loop length D. To maximize the voltage V(0) at the source end (which is where the measurements are also made in SELT), the product βD=nπ for n=0, 1, . . . , which, for traveling wave wavelength
      λ    =                  2        ⁢        π            β        ,results in
                    2        ⁢        D            λ        =    n    ,for n=0,1, . . . For a short-circuited lossless transmission line, the transmission line voltage V(x) is:V(x)=−2V0+e−βDi sin [β(x+D)]i  (7)The source end voltage V(0) is maximized when the product
      β    ⁢                  ⁢    D    =                    (                              2            ⁢            n                    +          1                )            ⁢      π        2  for n=0, 1, . . . , which result in
                    2        ⁢        D            λ        =                            2          ⁢          n                +        1            2        ,for n=0, 1 . . . These conclusions also apply to lossy transmission lines, as can be derived by those skilled in the art. Accordingly, one can solve for loop length D from these real and imaginary components of voltage V(0), at two or more peak frequencies.
FIG. 3 illustrates a simplified conventional FDR measurement scheme for SELT of transmission loop LOOP with load L. Sinusoid generator 32 applies a sinusoid sweep signal into loop LOOP through resistor load 35. The sum of the applied sinusoid sweep, and its reflected voltage from loop LOOP and load L, at node N is applied to one input of each of multipliers 34, 36. Multiplier 34 also receives the sinusoid sweep from sinusoid generator 32, while multiplier 36 receives a phase-shifted sinusoid (cosine) sweep from sinusoid generator 33. The outputs of multipliers 34, 36 are applied to amplifiers 37, 39, respectively, to produce the imaginary and real components of the voltage V(0,ω) at node N, at the source end of transmission loop LOOP. The frequency-domain voltage V(0,ω) at node N can be expressed as:
                              V          ⁡                      (                          0              ,              ω                        )                          =                                            V              t                        ⁡                          (              ω              )                                                                          H                r                            ⁡                              (                ω                )                                      ⁢                                          H                ht                            ⁡                              (                ω                )                                      ⁢                                          f                1                            (                              (                                                      Z                    L                                    ⁡                                      (                    ω                    )                                                  )                                                                        (        8        )            Once the product of Hr(ω))Hht(ω)ƒ1((ZL(ω)) is calibrated out, the peak frequencies of V(0,ω) and the wavelengths of the excitation signals from sinusoid generator 32 lead to the calculation of the loop length and reflection coefficient.
It has been observed, however, that the minimum loop length detectable by this conventional FDR approach is determined by the highest frequency of the input signal from sinusoid generator 32. In conventional DSL SELT, this highest tone is tone 63, from which the minimum detectable loop length is 1000 feet. Unfortunately, some loops may be shorter than this minimum length, and thus measurement of those loop lengths is not possible by FDR. In addition, because the loop length is determined by the ratio of two peak-frequencies, precision may be unacceptable if the peak frequencies are too close to one another.
By way of further background, a frequency domain reflectometry approach to characterizing a subscriber loop by estimations of a scattering parameter is described in Bostoen et al., “Estimation of the Transfer Function of a Subscriber Loop by Means of a One-Port Scattering Parameter Measurement at the Central Office”, IEEE J. Selected Areas in Communications, Vol. 20, No. 5 (June 2002), pp. 936–948.
Another conventional approach to SELT is time-domain reflectometry (TDR). According to this approach, the time delay Tp between an excitation signal and the reflected response depends upon the loop length D from the source to the location of the impedance mismatch:
                    D        =                                                            V                op                            ⁡                              (                f                )                                      ⁢                          T              p                                2                                    (        9        )            using the frequency-dependent velocity of propagation Vop(ƒ) along transmission loop LOOP. Unfortunately, loop attenuation is often so large that the reflection signal is too weak to be detected over long loops, because the amplitude of the reflection is dwarfed by the transient response of the forward going signal. In other words, the reflected signal depends on:v 0−(t)=v 0(t)−v0+(t)=v0(0,t,ZL)−v0(0,t,ZL0)  (10)where v0(0,t,ZL) is the measured waveform, and where v0(0,t,ZL0) is the waveform if the loop LOOP is properly terminated at load L, with no reflection.
FIG. 4 illustrates a conventional TDR instrument architecture. Pseudo-random sequence function 40 generates a pseudo-random digital sequence to be transmitted. The use of a pseudo-random sequence for TDR measurement provides the advantages of controlled power-spectrum density (PSD) within the requirements of the operative DSL standard, and also the ease of correlation with the reflection signal. Following filtering by transmit filter 42, digital-to-analog converter (DAC) 44 converts the filtered pseudo-random sequence to an analog signal that is transmitted over transmission loop LOOP via line interface 46. Line interface 46 also senses the voltage V(0) at the source end of loop LOOP, and applies this voltage to analog-to-digital converter (ADC) 47. After filtering by receive filter 48, the filtered received digital signal is applied to sequence correlator 49. Sequence correlator 49 performs digital matching of the filtered received signal against the output of pseudo-random sequence function 40, to determine the delay times Tp between points in the sequence of the output of pseudo-random sequence function 40 and the times at which these points in the sequence appear as reflected signals from loop LOOP. The delay times Tp are then processed to derive the loop parameters, as mentioned above.
By way of further background, a TDR methodology for loop qualification and characterization is described in Galli et al., “Loop Makeup Identification Via Single Ended Testing: Beyond Mere Loop Qualification”, IEEE J. Selected Areas in Communications, Vol. 20, No. 5 (June, 2002), pp. 923–935.
It has been observed, however, that conventional TDR is difficult to implement. Primarily, the weakness of the reflected signal often makes the correlation by sequence correlator 49 very difficult and imprecise. Short loop cases also require significant calibration to remove the transient response, especially with active termination involved in the transceiver and at short loop lengths.